Method and means for controlling a bridge circuit

ABSTRACT

A method and device for controlling a bridge circuit for providing current or power to a load. The bridge comprises two transistors including flywheel diodes and connected in series between positive and negative power supply rails for conducting current to or from the load under control of a control drive circuit. An LC-circuit is connected between the bridge circuit and the load. The bridge voltage (E) of the connection between the two transistors and the current (I) through the inductance of the LC-circuit are monitored and a firing pulse is supplied to one of the transistors. When the current (I) through the inductance exceeds a preset value (I&#39;), the transistors is turned OFF, whereupon the current of the inductance continues to flow another way through the flywheel diode of the opposite semiconductor member and consequently the bridge voltage (E) changes polarity a first time to the opposite rail polarity until the magnetic energy of the inductance has been terminated resulting in a second change of polarity of the bridge voltage (E) towards the first rail polarity. The change of the polarity of the bridge voltage (E) towards the first rail polarity is detected and another firing pulse is supplied at or shortly after said change.

BACKGROUND OF THE INVENTION

1. Field of Invention

This invention relates to a new switch control method for the well-knownbridge circuit, where two semiconductor switches and two flywheel diodesare connected in series in a bridge between a positive and a negativepower supply rail. The invention also relates to a bridge circuit forperforming said method.

2. Prior Art

A bridge circuit of the above-mentioned type usually comprises one ormore sets of components and one such set of components is normallycalled one "leg" of the bridge circuit and sometimes a "half-bridge".Very often two legs are used in a "full-bridge", with the load connectedbetween the two legs which are driven with opposite polarities. Inthree-phase systems, e.g. three-phase motor control circuits, three legsare used as is well-known. The present invention discloses one leg of abridge circuit, whereby this leg may be used as a building block in alltypes of systems with any number of legs.

The new method and the new switch circuit according to this inventionare especially adapted for using the POWER MOSFET transistor, where aflywheel diode is an integral part of the MOSFET transistor. However,the circuit may as well be used with any other type of semiconductor,such as bipolar transistors or thyristors, with external flywheeldiodes.

The POWER MOSFET transistor, in the following simply named transistor,has an integral "reverse diode", which can be used as a flywheel diodein the above-mentioned bridge connection. This is very favourable, sinceit minimizes the number of circuit components.

However, the use of said reverse diode as a fly-wheel diode is not freefrom problems. Such an application which may cause trouble is the pulsewidth modulated inverter for AC motor drives having inductive load. Theload current changes slowly, and may have the same polarity and beapproximately constant during multiple bridge output pulses.

Suppose in such a case, that the upper transistor in a bridge circuithas been turned ON and has supplied a positive output current to theload. When the transistor is turned OFF, the inductive load current mustfind a new way through the lower flywheel diode. But, in this type ofapplication, the flywheel current does not go to zero. The uppertransistor must be turned ON while the lower flywheel diode is stillconducting. Because of the internal transistor structure, details ofwhich are not discussed here, the lower transistor may beunintentionally turned ON when the upper transistor turns ON whichcauses a bridge short circuit with catrastrophic outcome. These arewell-known facts and are described in the literature, see e.g. RCA PowerMOSFET's Databook, pages 493-499 (1986), INTERNATIONAL IP. RECTIFIERPower MOSFET HEXFET® Databook, pages A-74-A-76 (1985), MOTOROLASEMICONDUCTORS TMOS® Power MOSFET Transistor Data, pages A-30-A-31(1985), and SIEMENS SIPMOS® Datenbuch 1984/85, pages 19-20.

Even with separate external flywheel diodes, the turn ON of thetransistor is critical. In the turn ON moment, the transistor has tosupply current to the output, and simultaneously supply reverse recoverycurrent to the opposite diode. If the turn ON is fast, as it should be,it will be necessary for the transistor to supply more than twice theload current during the turn ON moment.

Some transistor manufacturers have designed special transistor versionsfor such applications, e.g. SIEMENS with the FREDFET(Fast-Recovery-Epitaxial-Diode-Field-Effect-Transistor). The internalreverse diode has been modified to a fast recovery diode, while thereverse diode normally used has a relatively slow recovery. This measuremay partially overcome the problem, although it does not seem to removethe real source of problem.

SUMMARY OF THE INVENTION

The present invention discloses a new switch control method, whichsolves the above-mentioned problem in another way. The method may beused with all types of standard transistors. Of course the new switchmethod can be used with the abovementioned special transistors too.

The new switch method according to the present invention is based on theuse of an LC-filter connected between the bridge and the load. Thisfilter "isolates" the bridge from the load in such a way that the bridgecurrent can be allowed to reach zero in each switch cycle, although theload current is approximately constant during one or multiple switchcycles.

The output filter has the further advantage that it protects thetransistor junctions from external noise induced from the output cable.Further, the high frequency noise created by the bridge is isolated fromthe output cable.

In order to minimize the size of the LC-filter, the switch frequencyshould be much higher than the frequency of the load current. As anexample, the load current in AC motor control has a maximum frequency ofapproximately 100 Hz, while the switch frequency may be around 1-1000KHz.

Thus, the invention relates to a method for controlling a bridge circuitfor providing current or power to a load. The bridge circuit comprisesone or several legs, each comprising two semiconductor members connectedin series between positive and negative power supply rails. Eachsemiconductor member comprises a switchable member for conductingcurrent to or from the load in the forward direction of thesemiconductor member under control of a control drive circuit, and aflywheel diode for conducting current in the opposite direction. In thefollowing, the expression "bridge circuit" means one such leg.

According to the invention the method comprises the steps of connectingan LC-circuit between the bridge circuit and the load; monitoring thebridge voltage of the connection between the semiconductor members andthe current through the inductance of the LC-circuit; supplying a firingpulse to one of said switchable members of said semiconductor membersfor initiating the conduction thereof; terminating the conduction ofsaid switchable member when the current through the inductance exceeds apreset value, whereupon the current of the inductance continues to flowanother way through the flywheel diode of the opposite semiconductormember and consequently the bridge voltage changes polarity a first timeto the opposite rail polarity until the magnetic energy of theinductance has been terminated resulting in a second change of polarityof the bridge voltage towards the first rail polarity; sensing thechange of the polarity of the bridge voltage towards the first railpolarity or otherwise detecting that the bridge or inductance current iszero and supplying another firing pulse at or after said change.

Preferably the current through the inductance is monitored by measuringthe voltage difference over the inductance and calculating the currentby substantially integrating the voltage difference by an integratingcircuit. The preset value of the inductance current can be an externalsignal or can be obtained from a control amplifier based on a presetvalue of the required load voltage compared with the actual loadvoltage. The change of polarity towards the first rail polarity can besensed by a voltage comparator sensing when the bridge voltage differsfrom the second rail polarity by a value which is less than half thevoltage between the positive and negative rails. Preferably, the changeof polarity towards the first rail polarity is sensed by two voltagecomparators, one for each semiconductor member, whereby a window isgenerated in which window both comparators allow activation of thecorresponding semiconductor member.

The invention also relates to a device for performing theabove-mentioned method. The device comprises an LC-circuit connectedbetween the bridge circuit and the load; a monitor circuit formonitoring the bridge voltage of the connection between thesemiconductor members and the current through the inductance of theLC-circuit, said monitoring circuit comprising a first comparator fordetecting when the current through the inductance exceeds a preset valueand a second comparator for detecting when the bridge voltage changespolarity towards the corresponding rail polarity; the control and drivecircuit being adapted to provide a firing pulse to one of saidswitchable members of said semiconductor members for initiating theconduction thereof, the control and drive circuit being adapted toterminate the conduction of said switchable member when said firstcomparator determines that the current through the inductance exceedssaid preset value and to supply another firing pulse when said secondcomparator determines that the bridge voltage changes towards thecorresponding rail polarity.

Preferably, the device comprises a calculating circuit for caculatingthe current of the inductance from the voltage over the inductance bysubstantially integrating said voltage and a control amplifier forobtaining the preset value of the current of the inductance by comparingthe actual voltage over the load with a preset voltage and substantiallyintegrating the difference. Moreover, the device comprises a voltagecomparator for detecting when the bridge voltage differs from the secondrail polarity by a value which is less than half the voltage between thepositive and negative rails. A timer circuit may be adapted forinhibiting the conduction of the corresponding semiconductor member ifthe conduction thereof exceeds a predetermined time duration.

Thus, the turn ON of the relevant transistor takes place when the bridgecurrent is zero and the upper transistor turns ON when the bridgevoltage changes from minus to plus and the lower transistor turns ONwhen the bridge voltage changes from plus to minus. An advantage of themethod according to the invention is that the two transistors can neverbe ON simultaneously.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is described in further details below by means of apreferred embodiment of the invention and with reference to the appendeddrawings.

FIG. 1 is an oscilloscope view showing the behaviour of the bridgecircuit after a pulse is applied to one of the transistors in the bridgecircuit.

FIG. 2 is a time diagram showing the method according to the invention.

FIG. 3 is a circuit diagram of the preferred embodiment of theinvention.

FIG. 4 is a partial circuit diagram showing the calculating circuit andthe control amplifier of the preferred embodiment.

FIG. 5 is a circuit diagram of a comparator of the preferred embodiment.

FIG. 6 is a partial circuit diagram showing a timer of the preferredembodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

Turning firstly to FIG. 3, there is shown a circuit diagram of thepreferred embodiment according to the invention. The diagram shows abridge circuit 1 comprising two transistors 2, 3 connected to a load 4through an LC-filter 5, 6.

Several systems may be connected in parallel to the load. If they havethe same input signal, which is the preset value of current, they willshare the load current equally. The separate systems do not have tooperate in synchronism. Of course, several bridge transistors 2, 3 canbe connected in parallel.

The bridge voltage E on the connection point between the transistors 2,3 and the inductance 5 is shown by the diagram of FIG. 1. If the lowertransistor 3 is switched ON and then OFF, there is provided a pulseduring time period 7 and the bridge voltage E immediately goes to minus.When the pulse is terminated, the energy stored in the inductance mustbe released and forces the voltage E to the opposite polarity duringtime period 8, i.e. plus, and opens the flywheel diode of the uppertransistor 2. When the energy of the inductance 5 has been terminated,the flywheel diode of the upper transistor 2 is blocked. However, theblockage cannot take place immediately but a current in the oppositedirection must flow in order to charge or recover the flywheel diode(reverse recovery current). This opposite current induces a new butweaker magnetic field in the inductance 5 which in turn gives rise to anopening of the flywheel diode of the first-mentioned lower transistor 3as shown at time period 9. Then, the energy oscillates between theinductance 5 and the leakage capacitances, primarily the transistoroutput capacitances (not shown in FIG. 3) in the system as shown to theright during time period 10 until attenuated.

It is pointed out that this sequence of events takes place due to thefact that an LC-filter has been connected between the bridge and theload.

It appears from FIG. 1 that the oscilloscope diagram after thetermination of the pulse is a damped oscillation, the amplitude of whichbeing cut by the flywheel diodes a plus and minus the rail voltage.

The present invention uses this sequence of events in order to avoid orcircumvent the above-mentioned problem It is noted that it should becompletely safe to turn on the lower transistor during the time period9, while the lower flywheel diode is open and the flywheel diode of theopposite transistor is closed. The transistor then takes over theconduction from the flywheel diode and a new sequence follows.

In FIG. 2 there is shown a time diagram of the method according to theinvention. According to FIG. 2, the upper transistor is initiallyconducting (in FIG. 1 the lower transistor is conducting). The procedureis completely the same and there is no principal difference between themethod using the upper or the lower transistor. For positive outputcurrent the upper transistor is turned ON and the flywheel current flowsthrough the lower diode. For negative output current the lowertransistor is turned ON and the flywheel current flows through the upperdiode.

The upper diagram 2a shows the voltage E. It should be noted that thetime axis is not linear but is heavily expanded at the rise and falltimes. Diagram 2b shows the current I through the inductance 5.

The upper transistor is conducting and the voltage E is at plus duringthe time period I. The current I through the inductance 5 risesapproximately linearly (actually exponentially). When the current Ireaches a preset value I', a first comparator changes its state fromlogic "1" to logic "0" as shown in diagram 2c. The comparator controlsthe transistor and turns it OFF and the voltage decreases according tothe switching characteristics of the transistor as shown during timeperiod II. The voltage E passes below zero and reaches minus and theflywheel diode of the negative transistor opens at the start of the timeperiod III as explained with reference to FIG. 1. The energy of theinductance is terminated during time III and the current through theinductance is reversed in order to turn OFF the flywheel diode of thenegative transistor (reverse recovery current). When the flywheel diodeof the negative transistor is turned OFF, the voltage E rises duringtime period IV until the positive flywheel diode opens. This rise ofvoltage is sensed by a second comparator, which turns the positivetransistor ON for a further sequence during time period V. The output ofthe second comparator is shown in 2d. When the voltage E drops below-V', the comparator outputs a logical "0" as shown. When bothcomparators are outputting a logical "1", the positive transistor isturned ON. This procedure will be further explained below in connectionwith FIG. 3.

It is noted that the switch frequency is not constant but is high at lowloads and decreases with increasing loads.

In FIG. 3 there is shown a circuit diagram of a presently preferredembodiment of the invention. It is contemplated that this circuit can bemade at least partially as a custom designed integrated circuit orApplication Specific Integrated Circuit ASIC and thus, the circuitsolutions shown are only exemplary for explaining the invention.

The bridge circuit is shown to the right in FIG. 3 as explained above.Each transistor is driven by a drive circuit 11, 12. The drive circuitsare galvanically isolated from the remaining control circuitry byopto-couplers 13, 14 shown as a light emitting diode and a phototransistor.

Each opto-coupler is connected to the output of an AND gate 15, 16having two inputs. One of the inputs is connected to a first comparator19, which compares the actual current I through the inductance 5 with apreset value I'. When the actual current I is below the preset value I',the comparator 19 outputs a logical "1" to AND gate 15. When the presetcurrent I' is exceeded, the output from AND gate 15 is terminated asshown at time II in FIG. 2 at 2c. The transistor 2 is then turned OFF.The output from the first comparator 19 is inverted by inverter 20 andconnected to the lower AND gate 16.

The second input of each AND gate 15, 16 is connected to a secondcomparator 17, 18, corresponding to the second comparator mentionedabove. The comparator compares the voltage E with preset limit values-V' and +V' (V' is a positive value), respectively, as shown in FIG. 3.The positive comparator 17 (the upper) outputs a logical "1" when thevoltage E is above the limit -V' and the negative comparator 18 outputsa logical "1" when the voltage E is below the limit +V'. Thus, there isa window between +V' and -V' where both comparators outputs a logical"1". The purpose thereof will be explained below.

In FIG. 4 there is shown a calculating circuit and a control amplifierfor calculating the actual current I and the preset current I'. Thepreset value can be any value between plus and minus the maximum bridgecurrent. Consequently, the bridge circuit is short-circuit proof.Although not shown, it is possible to have an adjustable current limit.

The actual current (I) through the inductance can be measured byconventional means, or calculated by analog or digital circuits. If thedifference voltage (E-U) across the inductance is measured, the current(I) can be calculated according to the following formula:

    I=(E-U)/(R+s L)

where

I=inductance current

E=bridge voltage

U=output voltage to the load

R=resistance of the inductance

L=inductance

s=Laplace operator

The resistance of the inductance should be as low as possible, whichhowever creates a problem in the calculating circuit. The calculationformula can be seen as a "transfer function" for the calculatingcircuit. If R is small, the DC gain of the transfer function is veryhigh. Then any small but unavoidable offset in the measuring orcalculating circuits will be amplified to an unacceptable value.

It is possible to modify the calculating formula, by increasing R forexample 10 times, which decreases the DC gain and thus the influence ofpossible offset voltages, and adjusting L so that the calculated valuewill be approximately correct during the maximum pulse time. Thus, thecalculating error will be significant only for times longer than themaximum pulse time, which however is outside the operation area of thecalculating circuit and makes no harm.

A calculation circuit for calculating according to the above-mentionedformula is shown in FIG. 4. The voltage U is subtracted from the voltageE in a first OP-amplifier 21. The voltage U is first inverted and scaledby inverter 22 and then connected to the summing input of theOP-amplifier 21 at the negative input thereof. The voltage E isconnected to the same summing input through a scaling resistance 23. TheOP-amplifier is connected substantially as an integrator according tothe formula above and integrates the difference between voltages E andU. The result is the inverted inductance current (-I).

The preset current I' can be provided or generated in any conventionalmanner. The actual current (I) and the preset current (I') aretransferred to the summing input of a fast comparator 24 and the outputthereof is the difference between I and I'. The comparator has very highgain and thus, the output thereof is either high or low. The comparatoris further provided with a certain hysteres by the resistances 25 and26. The output from the comparator is inverted and buffered by theinverter 27 for providing I'-I which is the output signal provided bythe first comparator 19 in FIG. 3.

The circuitry of the entire system described above operates at a currentgenerator delivering output current to the load. It may be used in thisway with preset value of the current I' as the input control signal.However, it is often preferred to have a voltage generator and thepresent system is easily converted to a voltage generator. The outputvoltage U is measured and fed back to a conventional PI-regulator(Proportional Integrating), the output of which is the preset currentI'. Such an integrating regulator will automatically correct forDC-offsets in the current calculating circuit.

As shown in FIG. 4, the preset value I' of the current can be calculatedby a control amplifier from the actual output voltage to the load U anda preset output voltage U'. A voltage corresponding to the negativevalue of the preset output voltage U' is fed to the negative input of asecond OP-amplifier 28. The actual output voltage U is also fed to saidnegative input through a scaling resistor 29. The difference between theactual output voltage and preset output voltage U-U' is substantiallyintegrated by the OP-amplifier 28 and the output thereof corresponds tothe preset current I' and is fed to the negative input of the comparator24. The Zener diodes at the output of the OP-amplifier 28 limit themaximum preset current I'.

In FIG. 5, there is shown a current corresponding to the comparators 17and 18 in FIG. 3. The comparators are made in TTL-circuits and comprisesthree inverters 30, 31 and 32. The bridge voltage E is fed to a firstresistor 33 and 34 for each branch. A voltage corresponding to thedesired offset from the zero voltage is fed to a second resistor 35 and36 for each branch. Zero voltage is defined as midway between thepositive and the negative rail. The junction between the first and thesecond resistor is fed to the input of one inverter 30 or 32, the inputof which also being grounded by a third resistor 37 and 38. The resultis that when the bridge voltage drops so that E+V' is below zero, theupper inverter 31 outputs a logic "0" and closes the AND gate 15. Whenthe bridge voltage rises so that E-V' is above zero, the lower inverter32 outputs a logic "0" and closes the AND gate 16 (V' is a positivevalue). The further operation should be evident from the description inconnection with FIG. 3.

If the load impedance is too high (or the maximum output voltage is toolow) it is impossible to output the preset current to the load. In thatcase, the output voltage goes to maximum, and the correspondingtransistor remains constantly ON, which may be undesirable. The maximumpulse time can be limited by two retriggerable timers, one for eachbridge transistor.

In FIG. 6, there is shown a timer circuit 39 for terminating a pulse ifthe duration thereof exceeds a predetermined value whereby each AND gate15, 16 is provided with a third input, which is connected to the outputof a retriggerable timer, the two trigger inputs of which beingconnected to the two other inputs of the corresponding AND gate. If theinputs of the timer falls for a certain time duration dictated by anRC-circuit 40, the timer outputs a logic "0", which terminates thepulse. Otherwise, the timer outputs a logic "1".

It is evident that the current I is approximately triangular, with thepeak value twice the mean value. It is the mean value that flows to theexternal load. If the switch transistors are assumed to have constant ONresistance, the triangular current waveform causes a power loss in thebridge transistors during conduction that is 1/3 greater than it wouldhave been with rectangular current waveform.

It is also noted that the switching ON of the transistor takes placewhen the current in the bridge is zero (or very close to zero) This factminimizes the power dissipation of the transistor at the switch ONmoment, which is an essential advantage. In the present invention, thetransistors are always turned on in the right moment when the oppositeflywheel diode current is zero.

As mentioned above, there is a window, where both comparators 17, 18 areoutputting a logic "1". The reason for this is that when the power isturned on to the circuit, the oscillations must start. Dependent ontolerances of the components 21, 22, 24, 27, 28 etc, the output ofcomparator 24 will be either high or low at the initiation of thesystem. In either cases, the corresponding AND gate will be opened andthe oscillations will start, since the other input to both gates are atlogic "1".

It is noted in FIG. 2f that a lower firing pulse is generatedunintentionally. However, the duration of said firing pulse is veryshort. In a practical circuit, the influence of this firing pulse isslightly delayed and turns the lower transistor ON when the lowerflywheel diode is already conducting. However, when the preset currentis zero, the whole system will oscillate by help of these short pulses.When one transistor is turned OFF, the other transistor is turned ON,then the first transistor is again turned ON, etc. The pulse times willbe dependent on the delays in the various circuitry and the system willoscillate at a high maximum frequency, which in the present embodimentcan be around 300 KHz.

It is also noted that the diagram in FIG. 2 is idealized in that thetime delays in the different circuitry is not taken account of. However,such delays only improve the safety of the present circuit and have noharmful influence. As an example, study FIGS. 2c and 2d. If the risingedge in FIG. 2c occurs before the falling edge in FIG. 2d, a new veryshort upper firing pulse will unintentionally be generated. This is atype of electronic "race" that sometimes occurs in pulse circuits. Ofcourse, the designer must observe the possibility of such situations andtake appropriate counter measures. In this system, it is noted that itis only necessary that the transistor is turned on during the timeperiod 9 in FIG. 1. Thus, there is sufficient time for any delays.

It is also noted that the bridge system is able to operate in agenerative as well as a regenerative mode, feeding energy to the load orreceiving energy from the load back to the power supply rails.

Hereinabove, a preferred embodiment of the invention has been describedin details. However, it is clear to a skilled person that many detailsmay be modified without departing from the scope of the invention. Forexample, the sensing of the direction of change of the bridge voltage Ecan be determined by a differentiator instead of a comparator. It may bepossible to replace the sensing of the voltage change by sensing whenthe current in the bridge or inductance is zero and firing thetransistor shortly thereafter.

The present embodiment has a rail voltage of about 2×155 V and a maximumoutput current of 25 A (mean value 12.5 A). The inductance isapproximately 40 μH and the resistance of the inductance is about 10 mΩ.The capacitance is about 10 μF. It is preferred to connect thecapacitances to both rails, in which case the capacitors also filtersthe rail voltage. The magnetic field in the inductance is about 0.2 T.The frequencies are from about 1-1000 KHz, preferably from 5-300 KHz andchanges depending on the load. The above mentioned values are only givenas example and can be modified within large limits when improvedcomponents are manufactured, specifically the semiconductors and theinductance core. The load can be connected between the output and groundor in any other conventional manner to other legs in the bridge system.

An apparatus according to the present invention is useful for deliveringoutput voltage and current of both polarities. If only one polarity isrequired, as for example in conventional DC power supply units, theinvention can still be used. Then, all those components, which arenecessary for the opposite polarity, can be removed, resulting in asimpler circuit.

Further modifications should be obvious to a skilled person. Theinvention is only limited by the appended claims.

What is claimed is:
 1. A method for controlling a bridge circuit forproviding current or power to a load, said bridge circuit comprising oneor several legs, each comprising two semiconductor members connected inseries between positive and negative power supply rails, eachsemiconductor member comprising a switchable member for conductingcurrent to or from the load in the forward direction of thesemiconductor member under control of a control drive circuit, and aflywheel diode for conducting current in the opposite direction, themethod comprising the steps of:connecting an LC-circuit between thebridge circuit and the load; monitoring the bridge voltage (E) of theconnection between the semiconductor members and the current (I) throughthe inductance of the LC-circuit; supplying a firing pulse to one ofsaid switchable members of said semiconductor members for initiating theconduction thereof; terminating the conduction of said switchable memberwhen the current (I) through the inductance exceeds a preset value (I'),whereupon the current of the inductance continues to flow another waythrough the flywheel diode of the opposite semiconductor member andconsequently the bridge voltage (E) changes polarity a first time to theopposite rail polarity until the magnetic energy of the inductance hasbeen terminated resulting in a second change of polarity of the bridgevoltage (E) towards the first rail polarity, sensing the change of thepolarity of the bridge voltage (E) towards the first rail polarity orotherwise detecting that the bridge or inductance current is zero andsupplying another firing pulse at or after said change.
 2. A methodaccording to claim 1, further comprising the steps of monitoring thecurrent (I) through the inductance of measuring the voltage difference(E-U) over the inductance and substantially integrating the voltagedifference by an integrating circuit.
 3. A method according to claim 1,further comprising the step of obtaining the preset value of theinductance current (I') by a control amplifier based on a preset valueof the required load voltage (U') compared with the actual load voltage(U).
 4. A method according to claim 1, further comprising the step ofsensing of the change of polarity towards the first rail polarity by avoltage comparator sensing when the bridge voltage (E) differs from thesecond rail polarity by a value which is less than half the voltagebetween the positive and negative rails.
 5. A method according to claim4, further comprising the step of sensing the change of polarity towardsthe first rail polarity by two voltage comparators, one for eachsemiconductor member, whereby a window is generated, in which windowboth comparators allow activation of the corresponding semiconductormember.
 6. A means for controlling a bridge circuit for providingcurrent or power to a load, said bridge circuit comprising one orseveral legs, each comprising two semiconductor members connected inseries between positive and negative power supply rails, eachsemiconductor member comprising a switchable member for conductingcurrent to or from the load in the forward direction of thesemiconductor member under control of a control drive circuit, and aflywheel diode for conducting current in the opposite direction by saidmeans comprisingan LC-circuit connected between the bridge circuit andthe load; a monitor circuit for monitoring the bridge voltage (E) of theconnection between the two semiconductor members and the current (I)through the inductance of the LC-circuit, said monitoring circuitcomprising a first comparator for comparing when the current (I) throughthe inductance exceeds a present value (I') and a second comparator forcomparing when the bridge voltage (E) changes polarity towards thecorresponding rail polarity; a drive circuit being adapted to provide afiring pulse to one of said switchable members of said semi-conductormembers for initiating the conduction thereof; said control drivecircuit being adapted to terminate the conduction of said switchablemember when said first comparator determines that the current throughthe inductance exceeds said preset value (I') and to supply anotherfiring pulse when said second comparator determines that the bridgevoltage (E) changes towards the corresponding rail polarity.
 7. A meansaccording to claim 6, further comprising a calculating circuit forcalculating the inductance current (I) from the voltage (E-U) over theinductance by substantially integrating said voltage.
 8. A meansaccording to claim 6 further comprising a control amplifier forobtaining the preset value (I') of the current through the inductance bycomparing the actual voltage (U) over the load with a preset voltage(U') and substantially integrating the difference.
 9. A means accordingto claim 6, further comprising a voltage comparator for comparing whenthe bridge voltage (E) exceeds the second rail polarity by a value whichis less than half the voltage between the positive and negative rails.10. A means according to claim 6, further comprising a timer circuit forinhibiting the conduction of the corresponding semiconductor member ifthe conduction thereof exceeds a predetermined time duration.